I. Field of the Invention
The present invention relates to radio frequency signal generation and more particularly relates to oscillator circuits.
II. Description of the Related Art
Radio frequency (RF) signal generation entails deriving alternating current (AC) from direct current (DC) energy. One typical RF signal generator is an oscillator. An oscillator is typically comprised of an active device that is biased with DC energy and loaded to provide positive feedback to the active device. The active device may be, by way of example, a bipolar transistor (BJT), a field effect transistor (FET) or metal semiconductor field effect transistor (MESFET) or other device.
A critical factor in the performance of an oscillator is the phase noise of the resultant AC signal. Phase noise is created by frequency and phase jitter in the resultant AC signal. Phase noise may be observed in the resultant AC signal by examination of its spectrum in the frequency domain. Phase noise is typically measured in terms of a one sided spectral density of phase fluctuations, S.sub..phi. (f) of the desired AC signal at some frequency offset from the desired AC signal. For example, an exemplary phase noise requirement for an oscillator producing a frequency of 1.8 GigaHertz (GHz) is -110 dBc/Hz at 10 KHz, -130 dBc/Hz at 100 KHz, and -150 dBc/Hz at 1 MHz. The phase noise which is offset from the desired AC signal by a small amount (typically less than 10 KHz), is referred to as the close-in phase noise. The phase noise which is offset from the desired AC signal by a larger amount (typically greater than 10 kHz) is referred to as the far-out phase noise. FIG. 1 shows a typical spectrum of an AC signal generated by an oscillator operating at frequency fo. The phase noise at fo+.DELTA. represents the close-in phase noise while the phase noise at point fo+2.DELTA. represents the far-out phase noise.
Phase noise is created from a number of sources. One major contribution to phase noise that can be minimized by a thoughtful design is the bias dependent junction capacitance inherent in active devices. FIG. 2 is a block diagram showing the basic configuration of an oscillator comprising a bipolar transistor. In FIG. 2, Z1, Z2, and Z3 represent the equivalent impedances coupled to the active device. Z1, Z2, and Z3 may be created using lumped, distributed or active components. In FIG. 2, the junction capacitance is shown as three discrete capacitors, Ccb, Cce, and Cbe. The junction capacitance is inherent in the device. Depending on the topology of the oscillator design, either Z1, Z2, or Z3 may comprise the resonant tank which should be the major determining factor of the frequency of the resultant AC signal. Note that no matter which option is taken, one of the stray junction capacitors is in parallel with the resonant tank and affects the resultant resonant tank center frequency. Therefore, the stray junction capacitance influences the frequency of the resultant AC signal. If the value of the stray capacitance changes, the frequency of the resultant AC signal also changes. Small time dependent changes in junction capacitance result in changes in frequency that will create phase noise on the resultant AC signal.
The value of the stray capacitance is determined by a number of factors. The inherent characteristics of the device determine a steady state value of capacitance. In addition, the value of the inherent stray capacitance is a function of the temperature and DC bias point of the device. Thus, if the temperature of the device changes, the value of the stray capacitance changes. If the bias point changes, such as due to noise in the DC power supply, the value of the stray capacitance also changes. As noted above, if the value of the stray capacitance changes, the resultant AC signal also changes. The change creates phase noise in the resultant AC signal.
When designing a low noise oscillator, the effect of stray capacitance may be reduced by several means. Because the value of the stray capacitance is dependent on the bias point selected for steady state DC operation, the bias point should be chosen to be a point at which the variations and absolute value of the stray capacitance is relatively small. Also, the effect of the stray capacitance is dependent on the selection of a topology. Referring again to FIG. 2, typically the value of Cbe is larger than the value of Cce. Therefore, a common base configuration in which Z2 is the tank oscillator may be preferred. If we think in terms of lumped elements, Z2 may be modeled in terms of a parallel inductor/capacitor (LC) tank. With this model, Cce becomes a second parallel capacitor across the tank. In order to reduce the effect of Cce, the tank capacitor may be chosen relatively large in comparison with the Cce. However, at high frequencies, large capacitance values require very small values of inductance to produce the desired tank center frequency. For example, for a parallel inductor/capacitor tank, the tank resonant frequency is defined by: ##EQU1## where: .omega..sub.0 is the tank resonant frequency; C is the value of the tank capacitor: and
L is the value of the tank inductor. PA1 C represents the combined effects of capacitors 24, 26, 32, 34 and the stray capacitances of the active device 18. PA1 .DELTA.C is the change in capacitance. PA1 Ic is the AC collector current at .omega..sub.0 ; PA1 Vbe is the AC voltage across the base-emitter junction at .omega..sub.0 ; PA1 ABS(Gm) is the magnitude of Gm; and PA1 .theta. is the phase of Gm.
As the value of C is increased, the value of L must decrease in order to maintain the same resonant frequency. Thus, the value of C may not be increased without bound and must be limited with consideration for the realizable values of L. Also, in reality, large capacitors do not exhibit good capacitance qualities at high frequencies.
Phase noise may be reduced by the careful selection of an active device. For example, BJTs typically have better flicker noise (i.e. 1/f noise) characteristics than FET devices. However, certain BJT devices have lower operating frequencies than other devices such as GaAs FETs, and, therefore, may not be viable alternatives at very high frequencies. Also, bipolar devices are cheaper, more readily available and more likely to be available from multiple sources.
Some oscillator configurations allow the frequency of the resultant AC signal to change in response to a control signal. Typically an oscillator of this type is referred to as a voltage controlled oscillator (VCO.) FIG. 3 illustrates in a graphical form the output frequency and corresponding tuning gain factor as a function of control voltage of a typical oscillator. A curve 2, as reference to the left vertical axis, represents the frequency response of the oscillator circuit in MegaHertz (MHz) to control signal in Volts (V). A portion of the curve 2 corresponding to a control signal level between 4 and 10 Volts is approximately linear. A straight line 4 is printed over the curve 2 to aid the viewer in the detection of the linear region. If the curve 2 is mathematical differentiated with respect to the control signal, the result is the oscillator sensitivity or gain factor, Kv, typically having units of Hz/V. A curve 6 represents Kv as referenced to the right vertical axis. The curve 6 is typical of standard well-designed oscillators in that Kv has a relatively large region in which its value remains fairly constant.
A VCO may be incorporated into a phase lock loop (PLL) which may precisely tune the VCO to produce a desired output frequency. When a VCO is included in a PLL, Kv is one of the factors which determines the behavior of the loop including the stability of the loop, the resultant phase noise spectrum and the speed at which the loop responds to a change within the loop. If Kv is not linear, the loop characteristics are variable depending on the output frequency resulting in instability and possibly poor phase noise performance at some output frequencies. Therefore, when designing an oscillator for inclusion in a phase look loop, the flatness of Kv is an important factor both in the stability and noise performance of the loop.
With the increased prevalence high frequency consumer communication products, the need for low noise oscillators is also increasing. For example, in the United States, Personal Communication System (PCS) telephones which operate at 1.8 GHz are available to the general public in large quantities. The communication techniques used by these devices operate best when the resulting waveform is produced and received by an oscillator with low phase noise. However, due to the consumer nature of the product, the cost of such products must remain as low as possible in order for the market to enjoy a high level of penetration.
FIG. 4 is a schematic diagram of a typical prior art common base oscillator topology. The configuration of FIG. 4 is advantageous because the design minimizes the effects of the capacitance of the base-emitter stray capacitance, Cbe, through a capacitive step-down network. In FIG. 4, an active device 18 is DC biased by the resistors 12, 14 and 20 and an inductor 16. A capacitor 10 acts as a bypass capacitor and provides a short to ground (a common ground) to the base of the active device 18 at the frequency of oscillation. A capacitor 22 acts as a DC blocking capacitor between the tank circuit and the D.C. bias of the active device. The value of the capacitor 22 is chosen to be low impedance at the frequency of oscillation. The tank circuit of the oscillator is comprised of capacitors 24, 26 and 32, an inductor 28 and a varactor 34. The capacitors 24 and 26 act as a step down network. The impedance of the emitter-base junction is effectively increased by 1/n*n where n=(C24+C26)/C24. Therefore, the effect of the base-emitter parasitic capacitance, Cbe, in creating phase noise on the output frequency is reduced. In general, the capacitors 24 and 26 should be chosen as large as possible to reduce the effect of the collector-emitter parasitic capacitance, Cce, as well as the base-emitter parasitic capacitance, Cbe, on the total equivalent capacitance. Also, the capacitors 24 and 26 step down network will help to minimize any resistive loading on the resonant tank from R20. The varactor 34 acts as a variable capacitor. Varactor 34 can comprise back-to-back varactors 34a, as shown in FIGS. 6, 7, and 9, or a single varactor 34b, as shown in FIG. 8. The capacitance of the varactor 34 decreases as the DC voltage applied to it increases. As the capacitance of the varactor 34 changes, the frequency of oscillation of the circuit also changes. Thus, the frequency of oscillation of the oscillator of FIG. 4 is changed according to the voltage applied to it at the point Vtune. The oscillator is termed a voltage controlled oscillator (VCO).
In the ideal case, the oscillation frequency of the circuit shown in FIG. 4 is defined by the equation: ##EQU2## where: .omega. is the frequency of oscillation; L is the value of the inductor 28; and
Continuous small changes in the frequency produce phase noise. The changes in the oscillation frequency are related to the overall response of the circuit by the following equation: ##EQU3## where: .DELTA..omega..sub.0 is the change in output frequency; .DELTA.L is the change in inductance; and
One main contribution to the close-in phase noise which depends upon the design of the oscillator comes from the time dependent capacitance value, .DELTA.C, which is highly dependent on the junction capacitance of the active device. Note that in addition to the active device junction capacitance, the thermal and junction noise of the varactor 34 or other passive elements which are included in the resonant tank also contribute to the phase noise. However, in a well designed oscillator, the junction capacitance noise dominates the close-in phase noise while the far-out phase noise is dominated by the noise from these other sources.
The topology shown in FIG. 4 works well with bipolar devices which have a large signal transconductance greater than one and phase shift of exactly 180 degrees. The large signal transconductance is defined as follows: ##EQU4## where: Gm is the large signal transconductance of the active device which is generally a complex number;
In the ideal case for an oscillator, Gm has a phase shift of 180 degrees. Positive feedback occurs when the total feedback path is 360 degrees. In the ideal case, the active device provides 180 degrees of phase shift and the resonant tank provides an additional 180 degrees of phase shift to the total require phase shift of 360 degrees. However in typical active devices, the phase of Gm is a function of frequency and the value of Gm decreases from the ideal 180 degree phase shift at higher frequencies. Therefore, a typical oscillator will operate with the resonant tank providing more than 180 degrees of phase shift to make up for the phase of the Gm being less than 180 degrees.
True .omega..sub.0 is defined as the natural frequency of the resonant tank in the absence of the active device. At the true resonant frequency, the phase of the resonant tank is 180 degrees. At all other frequencies, the resonant tank frequency is equal to some other value. Thus, in the configuration shown in FIG. 4, if the active device contributes less than 180 degrees of the phase shift, the oscillator operates at a frequency offset from the true .omega..sub.0 of the resonant tank where the resonant tank provided enough additional phase to obtain true positive feedback in the oscillator (e.g., 360 degrees total phase shift).
FIG. 5 is a graphical representation of the frequency and phase characteristics of a parallel resonant tank circuit which can be used to illustrate the adverse effect of oscillating offset from true .omega..sub.0. A curve 40 is referenced to the left vertical axis and represents the impedance of a typical resonant tank as a function of frequency in MegaHertz (MHz). A curve 42 is referred to the right vertical axis and represents the phase characteristic of the typical resonant tank as a function of frequency in MHz. When a resonant tank oscillates at the true .omega..sub.0, the corresponding phase characteristic of the resonant tank are very linear. Due to the linear nature of the phase, the tuning gain of the oscillator, Kv, is more likely to be linear. Also, due to the linear nature of the phase, minor noise disturbances do not cause large non-linear phase noise. In comparison, if the resonant circuit oscillates below the true .omega..sub.0 at .omega..sub..theta., the corresponding phase characteristic is more likely to be non-linear.
For example, assume that the resonant circuit is forced to oscillate at .omega..sub..theta. as shown on FIG. 5. The corresponding phase at .omega..sub..theta. as shown at point 44 is extremely non-linear. As a result, the linearity of the tuning gain of the oscillator may be adversely effected, i.e. Kvco becomes more non-linear. As the tuning gain, Kv, of the oscillator increases, the phase noise due to the varactor and the control voltage applied to the varactor may also increase. Also, due to the non-linear nature of the phase, the same small disturbance which would cause a negligible phase distortion at .omega..sub..theta. creates a non-linear phase error which in turn causes large amounts of phase noise at .omega..sub..theta.. Many other factors can also contribute to nonlinear tuning gain curves such as the characteristics of the varactor and the inductor. By using the present invention, the negative effects of the characteristics of the inductor and varactor on tuning gain linearity are reduced.
One practical way to compensate for a device having a transconductance of less than 180 degrees at the desired frequency of oscillation is to decrease the value of capacitor 22 of FIG. 4. Although decreasing the value of the capacitor 22 may help offset a small shifts in .theta., the method has several deficiencies. For example, the small value of capacitor 22 corresponds to a large impedance at the oscillation frequency and, therefore, significantly reduces the gain in the feedback network as the collector voltage is now divided across the capacitors 22, 24 and 26. To eliminate the undesirable side effects due to decreasing the value of the capacitor 22, the present invention provides an improved means and method of compensation for the non-ideal phase shift of the active device.
Thus, it will be appreciated that there is a need in the technology for a means and method of providing a low noise, low cost oscillator. The present invention provides such an oscillator.